Welding or cutting power supply using phase shift double forward converter circuit (psdf)

ABSTRACT

A technique for dynamically adjusting an output voltage for a welding or cutting operation is provided. The technique allows for varying output voltage at the welding or cutting torch by manipulating the duty cycles of two forward converter circuits. The present disclosure provides methods and systems for increasing synchronized duty cycles in a pair of forward converter circuits in response to increasing output voltage demand then changing a phase shift between the duty cycles in response to further increases in output voltage demand. The present disclosure provides a controller designed to receive input signals and generate output pulse width modulation signals that control the duty cycle width and phase shift of the outputs of the forward converter circuits in response to these signals. Methods of accommodating for the time needed for the transformer core to reset via leading edge or lagging edge compensation are provided.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Non-Provisional Application of U.S. ProvisionalPatent Application No. 61/036,598, entitled “Welding or Cutting PowerSupply Using Phase Shift Double Forward Converter Circuit (PSDF)”, filedMar. 14, 2008, which is herein incorporated by reference.

BACKGROUND

The present disclosure relates generally to welding and cutting powersupplies, and more particularly, to a method and system for controllinga dual circuit inverter power supply.

Power supply circuits typically convert AC power to an output suitablefor welding or cutting operations. The output power is provided at anappropriate voltage and/or current level and may be controlled andregulated according to the process requirements. Many industrial weldingand cutting processes have dynamic load voltage and current requirementsthat cannot be met by a static power supply output. For instance,initiation of an arc, electrode characteristics, length of an activearc, operator technique, and so forth may all contribute to transientvoltage requirements. Oftentimes, these dynamic requirements, which areabove the average load conditions, are of short duration (˜1millisecond—a few seconds) and comprise only a small part of the overallwelding or cutting time. Accordingly, the power supply must be capableof providing both average and dynamic load requirements.

Single or double forward converter circuits are currently used tofulfill these dual requirements. The average load requirements typicallydetermine the thermal design of the power supply circuits, dictating thesize and rating of components such as transformers, heat sinks, powerdevices, cooling fans and so forth. However, for welding and cuttingpower supplies to accommodate short dynamic loads, components capable ofhandling the short but extreme requirements traditionally must bechosen. This generally results in a circuit with oversized components ora lack of efficiency when the power supply is operating at averageconditions. Accordingly, there exists a need for circuits that canbetter handle both static and dynamic load requirements without theinefficiencies of traditional designs.

BRIEF DESCRIPTION

The present disclosure provides a novel technique for dynamicallyadjusting an output voltage for a welding or cutting operation designedto respond to such needs. The technique allows for varying outputvoltage at the welding or cutting torch by manipulating the duty cyclesof two forward converter circuits. In particular, the present disclosureprovides methods and systems for increasing synchronized duty cycles ina pair of forward converter circuits in response to increasing outputvoltage demand then changing a phase shift between the duty cycles inresponse to further increases in output voltage demand. Presentembodiments provide a controller designed to receive input signals andgenerate output pulse width modulation signals that control the dutycycle width and phase shift of the outputs of the forward convertercircuits. Further, methods of accommodating for the time needed for thetransformer core to reset via leading edge or lagging edge compensationare provided.

DRAWINGS

These and other features, aspects, and advantages of the presentembodiments will become better understood when the following detaileddescription is read with reference to the accompanying drawings in whichlike characters represent like parts throughout the drawings, wherein:

FIG. 1 is a perspective view of an exemplary welding or plasma cuttingpower supply unit in accordance with aspects of the present disclosure;

FIG. 2 is a block diagram of the components of an exemplary welding orcutting power supply in accordance with aspects of the presentdisclosure;

FIG. 3 is a circuit diagram illustrating an exemplary embodiment of thepower supply comprising forward converter circuits in accordance withaspects of the present disclosure;

FIG. 4 is a diagrammatical representation of exemplary waveformsillustrating in phase duty cycles of two forward converter circuits inaccordance with aspects of the present disclosure;

FIG. 5 is a diagrammatical representation of exemplary waveformsillustrating out of phase duty cycles of two forward converter circuitsin accordance with aspects of the present disclosure;

FIG. 6 is a diagrammatical representation of exemplary waveformsillustrating in phase duty cycles of two forward converter circuitsoperating below an upper limit in accordance with aspects of the presentdisclosure;

FIG. 7 is a diagrammatical representation of exemplary waveformsillustrating in phase duty cycles of two forward converter circuitsoperating at an upper limit in accordance with aspects of the presentdisclosure;

FIG. 8 is a diagrammatical representation of exemplary waveformsillustrating out of phase duty cycles of two forward converter circuitsoperating at an upper limit in accordance with aspects of the presentdisclosure;

FIG. 9 is a diagrammatical representation of exemplary waveformsillustrating duty cycles of two forward converter circuits shifting outof phase via leading edge compensation in accordance with aspects of thepresent disclosure;

FIG. 10 is a diagrammatical representation of exemplary waveformsillustrating duty cycles of two forward converter circuits shifting outof phase via lagging edge compensation in accordance with aspects of thepresent disclosure;

FIG. 11 is a block diagram illustrating exemplary processing logic thatmay be used to control the pulse width modulation of the power supplyoutput in accordance with aspects of the present disclosure

FIG. 12 is a diagrammatical representation illustrating exemplarycurrent waveforms of two ideal forward converter circuits during powersupply operation in accordance with aspects of the present disclosure;

FIG. 13 is a diagrammatical representation illustrating exemplarycurrent waveforms of two non-ideal forward converter circuits duringpower supply operation in accordance with aspects of the presentdisclosure; and

FIG. 14 is a diagrammatical representation illustrating exemplarycontrol signals generated to correct for mismatched current levels inthe forward converter circuits in accordance with aspects of the presentdisclosure.

DETAILED DESCRIPTION

FIG. 1 illustrates an exemplary welding or plasma cutting power supplyunit 10 which powers, controls, and provides supplies to a welding orcutting operation in accordance with aspects of the present invention.The side of the power supply unit 10 that faces the user contains acontrol panel 12, through which the user may control the supply ofmaterials, such as power, gas flow, wire feed, and so forth, to awelding or cutting torch 14. A work lead clamp 16 typically connects toa workpiece to close the circuit between the torch 14, the work piece,and the supply unit 10, and to ensure proper current flow. It should benoted that in some embodiments, such as for stick welding operations,the torch 14 may be an electrode. The portability of the unit 10 dependson a set of wheels 18, which enable the user to move the power supplyunit 10 to the location of the weld.

Internal components of the power supply unit 10 convert power from awall outlet or other source of AC or DC voltage, such as a generator,battery or other source of power, to an output consistent with thevoltage, current, and/or power, requirements of a welding or cutting arcmaintained between the workpiece and the welding torch 14. FIG. 2illustrates an exemplary block diagram of components that may beincluded in the welding or plasma cutting power supply unit 10.Specifically, FIG. 2 illustrates a primary power supply 20 which, inoperation, outputs direct current (DC) to a welding or cutting powersupply 22 comprising a first converter circuit 24 and a second convertercircuit 26. The converter circuits 24, 26 operate to combine theirrespective outputs at a single node, which feeds into a filter inductor28 that supplies an output voltage 30 (i.e. V_out) for the welding orcutting operation. The welding or cutting arc 32 is supplied with awelding or cutting current 33 and is connected to ground 34. In oneembodiment, individual inductors may be utilized in place of the filterinductor 28. In other embodiments, the inductor 28 may have multiplewindings used to combine the outputs of the two converter circuits 24,26.

In one embodiment, the power supply 20 may be a DC source, such as abattery. In other embodiments, the power supply 20 may be a circuit thatrectifies incoming alternating current (AC), converting it to DC. In theexemplary block diagram shown in FIG. 2, each of the converter circuits24, 26 are connected to a single primary power supply 20. In otherembodiments, the circuits 24, 26 may be powered from separate powersupplies. In further embodiments, the circuits 24, 26 may be connectedin parallel or series to the primary power supply 20 at the capacitors36, 56 of the converter circuits 24, 26. In the embodiment where thecircuits 24, 26 are connected in series with a single primary powersupply 20, each converter circuit would only receive half the totalvoltage of the primary power supply 20, which may allow for the use oflower voltage components within the converter circuits 24, 26.

FIG. 3 is a circuit diagram illustrating one embodiment of the weldingor cutting power supply 22 comprising the two forward converter circuits24, 26 in accordance with aspects of present embodiments. As previouslydescribed, the primary power supply 20 provides DC power to the firstconverter circuit 24 and the second converter circuit 26. In the firstinverter circuit 24, a voltage is first supplied across a capacitor 36.A pair of power semiconductor switches 38, 40 then chops the DC voltageand supplies it to a transformer 42 on the side of a primary winding 44of the transformer 42. The transformer 42 transforms the chopped primaryvoltage to a secondary voltage, at a level suitable for a cutting orwelding arc, and supplies it to a secondary winding 46 of thetransformer 42. The secondary voltage is then rectified by rectifierdiodes 48, 50 and supplied to the filter inductor 28. A set of diodes52, 54 provide a free-wheeling path for the magnetizing current storedin the transformer 42 to flow when the pair of semiconductor switches38, 40 turn off, and thus reset the magnetic flux or energy stored inthe transformer core.

Similarly, in the second inverter circuit 26, a voltage is firstsupplied across a capacitor 56. A pair of power semiconductor switches58, 60 then chops the DC voltage and supplies it to a transformer 62 onthe side of a primary winding 64 of the transformer 62. The transformer62 transforms the chopped primary voltage to a secondary voltage andsupplies it to a secondary winding 66 of the transformer 62. Thesecondary voltage is then rectified by rectifier diodes 68, 70 andsupplied to the filter inductor 28. A set of diodes 72, 74 provide afree-wheeling path for the magnetizing current stored in the transformer62 to flow when the pair of semiconductor switches 58, 60 turn off, andthus reset the magnetic flux or energy stored in the transformer core.

The combined rectified secondary voltage is supplied to the welding orcutting power supply output 30 and a welding or cutting current 32 isoutput from the circuits 24, 26. In other embodiments, the forwardconverter circuits 24, 26 may include additional components or circuits,such as snubbers, voltage clamps, resonant “lossless” snubbers orclamps, gate drive circuits, pre-charge circuits, pre-regulatorcircuits, and so forth. Further, as previously noted, the forwardconverter circuits 24, 26 may be arranged in parallel or in series inaccordance with present embodiments, meaning that the capacitors 36, 56may be connected in series or in parallel. Additionally, in furtherembodiments, the output of the first converter circuit 24 and the outputof the second converter circuit 26 may be connected in series. In thisembodiment, a single ground would be configured to support both circuits24, 26, and the output of the diodes 48, 50 of the first convertercircuit 24 would couple with the output of the diodes 68, 70 of thesecond converter circuit 26 before entering the inductor 28.

FIG. 4 is a diagrammatical representation of exemplary waveformsillustrating two possible in phase duty cycles of the two forwardconverter circuits 24, 26 in accordance with aspects of presentembodiments. The semiconductor switches 38, 40 in the first convertercircuit 24 are switched on and off during a switching period 76,defining an active period 78 for the circuit that begins at a startingtime 80 and ends at a stopping time 82. The duty cycle or switch pulsewidth ratio for the first converter circuit 24 then becomes the activetime 78 divided by the switching period 76. The active period 78 isdefined by a leading edge 84 that begins the pulse and a lagging edge 86that ends the pulse. In one embodiment, the pulse width ratio of theswitches 38, 40 is limited to an upper limit of 50% of the switchingperiod 76 so the core of the transformer 42 can naturally reset eachcycle via the diodes 52, 54.

Similarly, the semiconductor switches 58, 60 in the second convertercircuit 26 are switched on and off during a switching period 76,defining an active period 78 for the circuit that begins at the startingtime 80 and ends at the stopping time 82. The duty cycle or switch pulsewidth ratio for the second converter circuit 26 then becomes the activetime 78 divided by the switching period 76. The active period 78 isdefined by a leading edge 88 that begins the pulse and a lagging edge 90that ends the pulse. In one embodiment, the pulse width ratio of theswitches 58, 60 is limited to an upper limit of 50% of the switchingperiod 76 so the core of the transformer 42 can naturally reset eachcycle via the diodes 72, 74. In the illustrated embodiment, the dutycycle for the first converter circuit 24 and the duty cycle for thesecond converter circuit 26 are equal and synchronous, dictating thatthe circuits are operating in phase. In such an arrangement, the weldingor cutting current 32 is split between the two converter circuits 24,26.

FIG. 5 is a diagrammatical representation of exemplary waveformsillustrating two possible out of phase duty cycles of the two forwardconverter circuits 24, 26 in accordance with aspects of presentembodiments. As previously described, the semiconductor switches 38, 40in the first converter circuit 24 are switched on and off during aswitching period 76, defining the active period 78 for the circuit thatbegins at the starting time 80 and ends at the stopping time 82. Theduty cycle or switch pulse width ratio for the first converter circuit24 then becomes the active time 78 divided by the switching period 76.The active period 78 is defined by the leading edge 84 that begins thepulse and the lagging edge 86 that ends the pulse. Similarly, thesemiconductor switches 58, 60 in the second converter circuit 26 areswitched on and off during a switching period 76, defining an activeperiod 78 for the circuit 26 as before. However, the active period 78now begins at a later starting time 92 and ends at a later stopping time94 that are distinct from the starting time 80 and stopping time 82 ofthe first circuit 24. The duty cycle or switch pulse width ratio for thesecond converter circuit 26 is still the active time 78 divided by theswitching period 76. However, the active period 78 is defined by a newleading edge 96 that begins the pulse and a new lagging edge 98 thatends the pulse.

In the illustrated embodiment, the duty cycle for the first convertercircuit 24 and the duty cycle for the second converter circuit 26 areequal but out of phase. In one embodiment, the pulse width ratios of thepair of switches 38, 40 in the first circuit and the pair of switches58, 60 in the second circuit are each limited to an upper limit of 50%of the switching period 76 so the core of the transformer 42 cannaturally reset each cycle. However, it may be possible to achieve aneffective duty cycle of approaching 100% for the combination of the twocircuits since the output from each of the respective transformerscombine at a common node. In the illustrated embodiment, a transformerturns ratio, meaning the secondary turns 46, 66 divided by the primaryturns 44, 64 may be less than the turns ratio of a transformer in anembodiment where the two converter circuits 24, 26 operate exclusivelyin phase. For an embodiment where the two circuits operate in phase, thepulse width ratio of the converter switches may be limited to 50%,meaning the combined outputs at 28 are in phase and can only supplyvoltage or power to the inductor 28 with a pulse width ratio of 50%. Inthe illustrated embodiment, with the two converter circuits 24, 26operating out of phase and each limited to 50%, the duty cycle of thecombined output at the inductor 28 can approach 100%. This means thetransformer turns ratio for each converter circuit 24 or 26, may bereduced by approximately 50%. As is well known in the art, thetransformer turns ratio is a function of the input voltage to theconverter circuit 24 or 26, the output voltage 30, and the operatingduty cycle to the input of the inductor 28.

FIGS. 6 through 8 illustrate exemplary waveforms representing outputsfrom the two forward converter circuits 24, 26 that may be producedduring a control method in accordance with present embodiments. In thisembodiment, the first converter circuit 24 may be manipulated to actlike a lagging circuit 24, and the second converter circuit 26 may bemanipulated to behave like a leading circuit 26. During this controlmethod, the two inverter circuits 24, 26 may operate in phase at lowduty cycles. As a control loop senses and reacts to a condition at thewelding or cutting torch, such as a welding or cutting arc strike or anincrease in transfer height from the torch to the workpiece, requiringgreater output voltage from the inverter circuits, the duty cycle mayincrease until it reaches an upper limit that may be defined by the timeneeded for the transformer core to reset. Beyond that point, the leadingcircuit 26 may be shifted out of phase and can continue shifting furtherout of phase until the two circuits 24, 26 are fully out of phase and anupper limit output voltage is being produced by the circuits 24, 26.

FIG. 6 illustrates an output of a first step of the control method. Thelagging circuit 24 output waveform is formed when the semiconductorswitches 38, 40 are switched on and off during a switching period 76,defining an active period 100 for the circuit that begins at a startingtime 102 and ends at a stopping time 104. The duty cycle or switch pulsewidth ratio for the circuit becomes the active period 100 divided by theswitching period 76. The active period 100 is defined by a leading edge106 that begins the pulse and a lagging edge 108 that ends the pulse.Similarly, the leading circuit 26 output waveform is formed when thesemiconductor switches 58, 60 are switched on and off during a switchingperiod 76, defining an active period 100 for the circuit that begins ata starting time 102 and ends at a stopping time 104. The duty cycle orswitch pulse width ratio for the circuit becomes the active period 100divided by the switching period 76. The active period 100 is defined bya leading edge 110 that begins the pulse and a lagging edge 112 thatends the pulse. The pulse width of each converter circuit is initiallyless than the 50% embodiment value. The duty cycle for the firstconverter circuit 24 and the duty cycle for the second converter circuit26 are equal and synchronous, dictating that the circuits are operatingin phase. The circuits would operate in this manner when the requiredoutput voltage was at or near the rated load, or less than rated load.

As the output voltage demand increases, the leading and lagging circuits24, 26 synchronously increase their respective active periods until theyreach an upper limit active period 114 that may be defined by the timeneeded for the transformer core to reset as illustrated in FIG. 7. Thelagging circuit 24 duty cycle has reached an upper limit active period114 that begins at the same starting time 102 and ends at a laterstopping time 116 with respect to FIG. 6. The pulse width 114 is definedby the same leading edge 106 and a later lagging edge 118 with respectto FIG. 6. Similarly, the leading circuit 26 duty cycle has reached anupper limit active period 114 that begins at the same starting time 102and ends at a later stopping time 120 with respect to FIG. 6. The pulsewidth 114 is defined by the same leading edge 110 and a later laggingedge 122 with respect to FIG. 6.

Once the duty cycles of the circuits have increased to their upper limit114 in response to a greater voltage demand, further increases in outputvoltage demand must be met via a phase shift of the leading circuit 26as illustrated in FIG. 8. The active behavior of the lagging circuit 24remains unchanged with respect to FIG. 7 while the leading circuit 26shifts out of phase to accommodate the further increase in voltagedemand. The pulse width of the active period 114 of the leading circuit26 remains unchanged, but the location of the leading edge 124 of theleading circuit pulse is shifted with respective to the leading edge ofthe lagging circuit 106 by an amount 126 dictated by the magnitude ofthe output voltage demand. The active period 114 of the leading circuit26 is defined by an earlier starting time 128, a later stopping time130, and a later lagging edge 132. The leading circuit 26 will continueto shift further out of phase as needed until the two circuits meet theoutput voltage demand or are fully out of phase, producing the upperlimit of their collective voltage output. This type of phase modulationmay occur when a transient high voltage requirement event occurs at thewelding or cutting arc, such as the initiation of a welding or cuttingarc. FIGS. 6-8 illustrate one embodiment in which three of the manypossible combinations of pulse width and phase shift amount are shown.In other embodiments, the two converter circuits continuously respond tooutput voltage demand by increasing and/or decreasing duty cycle and/orphase shift amount as needed.

The forward converter circuits 24, 26 use a natural transformer corereset mechanism where the magnetizing current can naturally flow throughthe free-wheeling diodes 52, 54 of the first converter circuit 24 andthe free-wheeling diodes of the second converter circuit 72, 74 duringan OFF interval of the switches of the first converter circuit 38, 40and the switches of the second converter circuit 58, 60, to allow thetransformer cores to reset, where the OFF interval refers to the portionof the switching period 76 that the circuit is not active. A method forallotting sufficient time for the transformer core to complete itsnatural reset cycle during phase shifting may need to be employed. Inone embodiment, the leading circuit 26 would skip a pulse when the phaseshift value was increasing or decreasing to reset to the proper phaseshift. Outputs for other possible embodiments of such a method areillustrated by the exemplary waveforms in FIGS. 9 and 10.

FIG. 9 is a diagrammatical representation of exemplary waveformsillustrating duty cycles of the two forward converter circuits 24, 26shifting out of phase and compensating for the necessary transformercore reset time via leading edge compensation. In this embodiment, thelagging circuit 24 proceeds as previously described with respect to FIG.6. The semiconductor switches 38, 40 are switched on and off, definingan active period 114 that begins at a starting time 102 and ends at astopping time 116. The active period 114 is defined by a leading edge106 that begins the pulse and a lagging edge 118 that ends the pulse,defining a duty cycle operating at an upper limit. However, the leadingcircuit 26, which was operating at an upper limit of its duty cycle 114and an initial phase shift 126, must again shift to a new phase shiftvalue 134 to accommodate a further increase in voltage output demand atthe welding or cutting torch. To allow sufficient time for thetransformer core to reset, some action must be taken to reduce the pulsewidth of the next pulse during the time that the phase shift isincreasing. The embodiment in FIG. 9 shows a single reduced pulse width136 formed by a new leading edge 138 and a new lagging edge 140, whichreflect a delay in the desired leading edge 142 to allow for a full offperiod 144 for the transformer core to reset. Succeeding pulses wouldreturn back to the original upper limit pulse width 114 at the new phaseshift value 134.

FIG. 10 is a diagrammatical representation of exemplary waveformsillustrating duty cycles of two forward converter circuits shifting outof phase and compensating for the necessary transformer core reset timevia lagging edge compensation. In this embodiment, the lagging circuit24 proceeds as previously described with respect to FIG. 6. Thesemiconductor switches 38, 40 are switched on and off, defining anactive period 114 that begins at a starting time 102 and ends at astopping time 116. The active period 114 is defined by a leading edge102 that begins the pulse and a lagging edge 116 that ends the pulse,defining a duty cycle operating at an upper limit. However, the leadingcircuit 26, which was operating at an upper limit of its duty cycle 114and an initial phase shift 126, must again shift to an increased phaseshift value 146 to accommodate a further increase in voltage outputdemand at the welding or cutting torch. As previously stated, to allowsufficient time for the transformer core to reset, some action must betaken to reduce the pulse width of the next pulse during the time thatthe phase shift is increasing. The embodiment in FIG. 10 shows a singlereduced pulse width 148 formed by a new leading edge 150 and a newlagging edge 152. In this embodiment, the desired start time of theleading edge 150 according to the new phase shift value 146 is notaltered. Instead, an early lagging edge 152 is initiated to accommodatethe need for adequate transformer core reset time. Even though thetransformer core was not fully reset before the onset of the next pulse,as indicated by the shortened inactive period 154, it was driven by apulse of reduced width due to the early onset of the lagging edge 152 sothat the peak flux in the transformer core at the end of the reducedpulse was no greater than it would be after a normal cycle when fullreset was allowed to occur. Succeeding pulses would return back to theoriginal upper limit pulse width 114 at the increased phase shift value146.

FIG. 11 is a block diagram illustrating exemplary processing logic thatmay be used to control the pulse width modulation of a power supplyoutput. In accordance with aspects of present embodiments, in theillustrated embodiment, a controller 156 comprising a regulator 158 anda processor 160 controls switching of forward converter circuits (e.g.forward converter circuits 24, 26) to achieve the desired voltage and/orcurrent output at the welding or cutting torch 14. In one embodiment,the controller 156 may use feedback information to prevent the convertercircuits 24, 26 from operating continuously in a full or partially phaseshifted manner. Additionally, the controller 156 may take other actionsto protect or prolong the life of the converter circuits 24, 26, such asreducing the output load current when operating beyond a certain timelimit in a phase shift mode. These actions could be taken to prevent thetwo inverter circuits 24, 26 from operating for an excessive amount oftime in a phase shifted mode at high currents, in the event of someabnormal dynamic load at the torch 14. The controller 156 may alsoprevent the converter circuits 24, 26 from operating in a phase shiftedmode if the output current is greater than some defined level.

The regulator 158 is configured to receive multiple inputs regarding thedesired and actual output voltage, current, power, and so forth. Forinstance, the regulator 158 may receive feedback from a current sensor162 and/or voltage sensor 164 at the welding or cutting torch 14.Additionally, the regulator 158 may receive a manual input 166 from pushbuttons, a user interface, voice command, and so forth, regarding adesired set point or output. The regulator 158 then generates an outputcontrol signal 168 based on its inputs. In one embodiment, the regulator158 may include an error amplifier and compensation network and may beimplemented with discrete circuits or software algorithms within theprocessor 160 or controller 156. The processor 160 may receive auxiliaryfeedback or input signals 170 such as temperature feedback, monitoringsignals, control signals, and so forth. The control signal 168 is alsoreceived by the processor 160 and is used to set the required pulsewidth modulation (PWM) signals, PWM1 172 and PWM2 174. The individualPWM signals 172, 174 may include pulse width and phase shift values asdictated by the output demand of the system. The two PWM outputs 172,174 are connected to two gate drive circuits 176, 178 to provide thenecessary gate drive signals to drive the switching of the firstconverter circuit 180 and the switching of the second converter circuit182. In certain embodiments, the processor 160 may output additionalauxiliary signals 184, such as analog or digital outputs for monitoringand control of aspects of the welding or cutting power supply. Auxiliarysignals 184 may include fan control signals, pre-charge relay signals,timing signals for other power circuits such as a pre-regulator circuit,and so forth.

As a control method in accordance with present embodiments isimplemented, the active states and phase shifts of the convertercircuits 24, 26 may be altered, and current flow waveforms may begenerated. For a particular phase shift and duty cycle, the exemplarycurrent waveforms shown in FIG. 12 may be generated for an ideal circuit(i.e. a circuit with optimal transformer coupling and so forth). In oneembodiment, when either the leading converter circuit 26 or the laggingconverter circuit 24 is exclusively on, the respective semiconductorswitches, either 38 and 40 or 58 and 60, will carry the full peakcurrent 186, which is dictated by the welding load or output current ofthe combined circuit outputs and the transformer turns ratios, aspreviously described. The full peak current 186 is effectively theoutput current multiplied by the transformer turns ratio. During thetime when both converter circuits 24, 26 are active, the load outputcurrent will split between the two circuits such that each carriesapproximately half of the peak current 188. The output currentmultiplied by the turns ratio of the transformer 28 results in eachconverter circuit 24 or 26 carrying one half of the peak current. Duringthe time when only one converter circuit is active, the other convertercircuit will carry no current 190. When the two converter circuits arefully in phase, such as would occur when operating at a rated loadoperating point, the currents in the primary switches 38, 40, 58, 60will be at approximately half of the peak current for the full activeportion of the cycle for each inverter circuit.

For a particular phase shift and duty cycle, the exemplary currentwaveforms shown in FIG. 13 may be generated for a non-ideal circuit(i.e. a circuit with real components). In one embodiment, when eitherthe leading converter circuit 26 or the lagging converter circuit 24 isexclusively on, the respective semiconductor switches, either 38 and 40or 58 and 60, will carry the full peak current 186, which is dictated bythe welding load or output current 33 of the combined circuit outputsand the transformer turns ratios, as previously described. During thetime when both converter circuits 24, 26 are active, the load outputcurrent 192 in the lagging circuit 24 is lower than ideally expected(˜50% of the peak current) with respect to FIG. 12. Since the total peakcurrent remains the same with respect to FIG. 12, the leading circuit 26carries a current 194 that is higher than ideally expected (˜50% of thepeak current) to compensate for the decrease in current in the laggingcircuit 24. During the time when only one converter circuit is active,the other converter circuit will carry no current 190.

This non-ideal sharing occurs because the voltage on the secondarywinding 46 of the transformer 42 of the lagging circuit 24 isapproximately equal to the voltage on the secondary winding 66 of thetransformer 62 that the leading circuit 26 is providing. This voltagereflects to the primary winding 44 of the transformer 42 of the laggingcircuit 24 and is nearly equal to the voltage available to drive thelagging circuit 24. Accordingly, there exists very little forcingvoltage to overcome the leakage inductance in the transformer 42 of thelagging circuit 24 until the leading circuit 26 enters an inactivestate, and the secondary voltage approaches zero. At that point, thefull voltage is available to ramp up the current in the primary winding44 of the transformer 42 of the lagging circuit 24, and it quicklyadjusts to carry the full peak current.

For transient, dynamic load conditions, the described mismatch in thecurrent carried by the converter circuits 24, 26 may be insignificant.However, in response to loads of longer durations or for embodiments inwhich the circuits 24, 26 are arranged in series, it may be desirable tobalance the currents carried by the inverter circuits 24, 26 todistribute power losses and heat generated during operation. Themismatch in average current carried by the converter circuits 24, 26 maycause a mismatch in voltages between the converter circuits 24, 26,causing unequal splitting of the total input voltage from the primarypower supply 20 when the circuits 24, 26 are connected in a seriesarrangement. This mismatch in voltage for the series arrangement maycause excess voltage stress on the components of the converter circuits24, 26. In one embodiment, this voltage mismatch may be compensated forby splitting the overlap time between the two circuits 24, 26 when theyare operating in a phase shifted mode. The duty cycles of the leadingand/or lagging circuit 24 and/or 26 may be adjusted such that theleading circuit 26 does not carry significantly more average currentthan the lagging circuit 24. In one embodiment, this adjustment wouldcomprise alternating which converter circuit 24 or 26 is the leadingcircuit during operation such that the circuit that carries a greaterportion of the peak current alternates.

In another embodiment as illustrated in FIG. 14, information regardingwhether the center point of the primary power supply input voltages foreach of the circuits 24, 26 is greater or lesser than half of the totalsupplied primary power supply voltage may be acquired and used tocompensate for the mismatch in current carried by the circuits 24, 26.In one embodiment, a comparator circuit may indicate this informationand provide it to the controller 156 by a single digital input line.This line may indicate whether the center point is too high or too lowfor equal current sharing between the two circuits 24, 26. Thecontroller 156 may then signal a duty cycle change in the circuits 24,26 to correct for the mismatch. FIG. 14 illustrates two sets ofwaveforms showing the control signals for two different time points. Thecontrolling variable 196 for the voltage output is identical for the twosets of waveforms, indicating that the desired voltage output has beenmaintained. The waveforms illustrate a larger leading duty cycle 198 anda reduced leading duty cycle 200, indicating that the active time of thelead converter circuit has been modified as required to correct for theunequal split of the primary power supply output voltage. The duty cycleof the lagging circuit remains the same 202. Effectively, the duty cycleof the leading converter 26 may be modulated in response to sensedvariations in the center point of the primary power supply outputvoltage. Additionally, it may be necessary or desirable under certainload conditions to modify or modulate the duty cycle of the laggingcircuit 24, as a further means of reducing a mismatch in the voltagesbetween the converter circuits 24, 26. Other embodiments may providemultiple signals or other types of signals, such as analog feedbacksignals indicating the magnitude of the voltages applied to the twoconverter circuits, to the controller 156. These signals may be used inplace of or in addition to the single digital input line described.

While only certain features of the present disclosure have beenillustrated and described herein, many modifications and changes willoccur to those skilled in the art. It is, therefore, to be understoodthat the appended claims are intended to cover all such modificationsand changes as fall within the true spirit of the present disclosure.

1. A method of dynamically adjusting an output voltage for a welding orcutting operation, comprising: increasing synchronized duty cycles in apair of forward converter circuits to accommodate an increase in avoltage requirement; and changing a phase shift between the duty cyclesupon reaching an upper limit for each of the duty cycles to accommodatea further increase in the voltage requirement.
 2. The method of claim 1,wherein the upper limit for each of the duty cycles is based on timerequired for a transformer core in each of the respective forwardconverter circuits to reset.
 3. The method of claim 1, wherein changingthe phase shift comprises delaying a leading edge of a pulse in aswitching period to reduce the duty cycle of a one of the pair offorward converter circuits.
 4. The method of claim 1, wherein changingthe phase shift comprises initiating an early lagging edge of a pulse ina switching period to reduce the duty cycle of a one of the pair offorward converter circuits.
 5. The method of claim 1, wherein changingthe phase shift comprises skipping a pulse in a switching period of aone of the pair of forward converter circuits.
 6. The method of claim 1,wherein changing the phase shift comprises increasing the duty cycles toa maximum limit, shifting outputs of the pair of forward convertercircuits fully out of phase, and reducing pulse widths in the switchingperiod for each of the forward converter circuits.
 7. The method ofclaim 1, comprising preventing saturation of a transformer in one of thepair of forward converter circuits by monitoring voltage and/or currentin the transformer and disabling initiation of a new pulse when thevoltage and/or current indicates that the transformer has not beenreset.
 8. The method of claim 1, comprising flowing current through thepair of forward converter circuits in parallel or in series.
 9. Awelding or cutting system, comprising: a pair of forward convertercircuits capable of coordinating to accommodate an increase in a voltagerequirement; and a controller, comprising: a regulator capable ofreceiving at least one input related to a desired combined output forthe pair of forward converter circuits, and capable of generating acontrol signal; a processor capable of receiving the control signal anddriving switches in the pair of forward converter circuits to facilitateshifting of duty cycles of the pair of forward converter circuits. 10.The welding or cutting system of claim 9, wherein the processor isconfigured to receive a temperature signal and adjust the switches toaccommodate a desired temperature setting.
 11. The welding or cuttingsystem of claim 9, wherein the processor is configured to output andadditional control signal for an auxiliary device.
 12. The welding orcutting system of claim 11, wherein the auxiliary device comprises afan.
 13. The welding or cutting system of claim 9, wherein regulatorcapable of receiving a measured value for an actual output from anoutput sensor.
 14. The welding or cutting system of claim 9, comprisinga phase shift duration controller configured to limit continuousoperation in a phase shifted manner.
 15. (canceled)
 16. The welding orcutting system of claim 10, wherein the pair of forward convertercircuits are arranged in parallel or in series.
 17. A method ofaccommodating a voltage requirement of a welder or plasma cutter,comprising: controlling current through a first forward convertercircuit; controlling current through a second forward converter circuit,wherein the first and second forward converter circuits cooperativelysupply a voltage output; and shifting a phase of a first currentwaveform through the first forward converter circuit relative to asecond current waveform through the second forward converter circuitbased on the voltage requirement.
 18. The method of claim 17, whereinthe current through the first and second forward converter circuits isapproximately equal when the first and second forward converter circuitsare operating fully in phase.
 19. The method of claim 17, comprisingpassing substantially all available current through the first forwardconverter circuit when the first and second forward converter circuitsare operating completely out of phase.
 20. The method of claim 17,comprising balancing voltage supplied to the first and second forwardconverter circuits when the first and second forward converter circuitsare in series.
 21. The welding or cutting system of claim 9, wherein theprocessor is capable of facilitating an increase in the duty cycles to amaximum limit by shifting outputs of the pair of forward convertercircuits fully out of phase.